System and method for frequency translation with harmonic suppression using mixer stages

ABSTRACT

A method for generating phase signals includes triggering a phase register to output a binary number stored in the phase register, wherein the phase register is triggered based at least in part on a voltage signal provided by a voltage controlled oscillator. The method also includes providing an input signal to a decoder, wherein the input signal is based at least in part on the binary number output by the phase register and the decoder is operable to generate phase signals in response to the input signals. The method further includes incrementing the binary number stored in the phase register and repeating the triggering and providing steps after the binary number is incremented.

RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.11/622,688 filed Jan. 12, 2007 entitled “System and Method for FrequencyTranslation with Harmonic Suppression Using Mixer Stages,” now U.S. Pat.No. 7,610,032.

Application Ser. No. 11/622,688 is a continuation of application Ser.No. 10/769,398, filed Jan. 30, 2004 entitled “System and Method forFrequency Translation with Harmonic Suppression Using Mixer Stages,” nowU.S. Pat. No. 7,164,899

Application Ser. No. 10/769,398 is a continuation-in-part of U.S.application Ser. No. 10/663,824 filed Sep. 16, 2003 entitled, “Systemand Method for Frequency Translation with Harmonic Suppression UsingMixer Stages,” now U.S. Pat. No. 7,190,943.

TECHNICAL FIELD OF THE INVENTION

This invention relates to circuits and more particularly to frequencytranslation with harmonic suppression using mixer stages.

BACKGROUND OF THE INVENTION

Mixers are the circuit blocks of a communication system that performfrequency translation of the carrier signals. Mixers are therefore usedto frequency translate a desired radio frequency (RF) signal from abroadband signal to an intermediate frequency (IF) signal. Ideally, afrequency translation receiver, such as a direct down-conversionreceiver, using a mixer multiplies the RF signal of interest by a puresine wave, known as the local oscillator (LO) signal. This idealmultiplication produces signals only at the sum and difference of the RFand LO frequencies. With low pass filtering of the multiplier output,the receiver responds only to the signals at the frequency of interest,i.e. RF signals in a small band centered about the LO frequency.

Unfortunately, ideal multipliers are not practical for a variety ofreasons. Currently, standard integrated circuit (IC) practice is toimplement the mixing process with a Gilbert cell. A Gilbert cellessentially multiplies the RF signal by a square wave rather than anideal sine-wave. Because of the odd harmonics of a square wave, areceiver utilizing a Gilbert cell mixer responds to RF signals at eachof the odd harmonics of the LO. Response to the first harmonic isstrongest; higher harmonics have a weaker, but significant, response.For example, the third and fifth harmonic responses are 9.5 and 14 dBbelow the first harmonic, respectively. Prior approaches address theharmonic problem by placing a pre-selection filter before the mixer. Forwide band applications, the filter must be tunable. The filter passesonly the RF signal of interest and greatly attenuates its harmonics.Since harmonics of the RF signal never reach the mixer, the receiverresponds only to the signal of interest. Unfortunately, a suitablepre-select filter is difficult or impossible to implement with currentIC technology.

SUMMARY OF THE INVENTION

In accordance with the present invention, the disadvantages and problemsassociated with prior frequency translation circuits have beensubstantially reduced or eliminated.

In accordance with one embodiment of the present invention, a method forgenerating phase signals includes triggering a phase register to outputa binary number stored in the phase register, wherein the phase registeris triggered based at least in part on a voltage signal provided by avoltage controlled oscillator. The method also includes providing aninput signal to a decoder, wherein the input signal is based at least inpart on the binary number output by the phase register and the decoderis operable to generate phase signals in response to the input signals.The method further includes incrementing the binary number stored in thephase register and repeating the triggering and providing steps afterthe binary number is incremented.

The following technical advantages may be achieved by some, none, or allof the embodiments of the present invention. Technical advantages of thefrequency translation circuit include suppression of the harmonicsassociated with a fundamental frequency for a signal of interest. Theseand other advantages, features, and objects of the present inventionwill be more readily understood in view of the following detaileddescription and the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention and itsadvantages, reference is now made to the following description, taken inconjunction with the accompanying drawings, in which:

FIG. 1 illustrates one embodiment of a circuit for suppressing theharmonics of a radio frequency signal according to the teachings of thepresent invention;

FIG. 2 illustrates phase signals to be used in the circuit of FIG. 1;

FIG. 3 illustrates an intermediate frequency signal generated by thecircuit of FIG. 1;

FIG. 4 illustrates a table with example data for the operation of thecircuit in FIG. 1;

FIG. 5 illustrates one embodiment of a circuit that uses a switchingcircuit and a plurality of mixer stages to suppress the harmonics of aradio frequency signal;

FIG. 6 illustrates one embodiment of a mixer stage used in the circuitof FIG. 5;

FIG. 7 illustrates one embodiment of a phase generation circuit used inthe mixer stage of FIG. 6;

FIG. 8 illustrates one embodiment of phase signals generated by thephase generation circuit of FIG. 7;

FIG. 9 illustrates another embodiment of a mixer stage used in thecircuit of FIG. 5;

FIG. 10 illustrates one embodiment of a phase generation circuit used inthe mixer stage of FIG. 9;

FIG. 11 illustrates one embodiment of phase signals generated by thephase generation circuit of FIG. 10;

FIG. 12 illustrates yet another embodiment of a mixer stage used in thecircuit of FIG. 5;

FIG. 13 illustrates one embodiment of a phase generation circuit used inthe mixer stage of FIG. 12;

FIG. 14 illustrates one embodiment of phase signals generated by thephase generation circuit of FIG. 13; and

FIG. 15 illustrates one embodiment of a phase generation circuit thatmay be used with various embodiments of the mixer stages of the circuitof FIG. 5;

FIG. 16 illustrates a truth table corresponding to one embodiment of ann-to-N decoder in the phase generation circuit of FIG. 15;

FIG. 17 illustrates one embodiment of a phase generation circuit thatmay be used with various embodiments of the mixer stages of the circuitof FIG. 5; and

FIG. 18 illustrates one embodiment of phase signals generated by thecircuit of FIG. 17.

DETAILED DESCRIPTION OF EXAMPLE EMBODIMENTS

FIG. 1 illustrates one embodiment of a circuit 10 for suppressing theharmonics of a radio frequency (RF) signal 12 to be frequency translated(e.g., down-converted or up-converted) to an intermediate frequency (IF)signal 14. Circuit 10 comprises an array of N mixers 16 to approximatethe multiplication of RF signal 12 by an ideal sine-wave. Each mixer 16multiplies the RF signal 12 by a phase signal 18 having a magnitude of,for example, either plus or minus one. The RF signal 12 is weightedaccording to a weighting factor (e.g., multiplied by w_(i)) before theinput to each mixer 16. All mixer outputs 22 are summed by a summingcircuit 24 to generate the IF signal 14.

In a television system, signals representing individual channels areassigned to specific frequencies in a defined frequency band. Forexample, in the United States, television signals are generallytransmitted in a band from 48 MHz to 852 MHz. In such televisionsystems, RF signal 12 comprises a radio frequency signal in the bandfrom 48 MHz to 852 MHz. The phase signal 18 of each of the mixers 16 isa square wave generated by phase generation circuit 26 at the frequencyof interest (e.g., frequency of the signal of interest). As an example,FIG. 2 illustrates the phase relationship between the phase signals 18for an array of four mixers 16. These staggered phase signals 18 can begenerated by digital logic clocked by a voltage controlled oscillator(VCO) 28 that runs at a multiple (e.g., 2N) of the frequency of interestand that provides a VCO signal 30 to phase generation circuit 26. Ingeneral, IF signal 14 comprises a combination of RF signal 12 and phasesignals 18. If the RF signal 12 is up-converted, thenf_(IF)=f_(RF)+f_(LO). If the RF signal 12 is down-converted, thenf_(IF)=f_(RF)−f_(LO). IF signal 14 may include a real part, I, and animaginary part, Q, as discussed in greater detail below.

Mixers 16 comprise any suitable device or circuitry that multiplies anRF signal 12 with a phase signal 18 to generate an IF signal 14, or atleast an output 22 that comprises a portion of IF signal 14. Mixers maybe formed using suitable Bipolar, CMOS and BiCMOS transistortechnologies. In a particular embodiment, mixers 16 comprisedouble-balanced quad mixers, which are often referred to as Gilbert cellmixers. In such a mixer, for example, an input signal voltage isconverted to a current using an emitter coupled pair. The current isthen switched back and forth by a quad switch to produce frequencyconversion of the input signal. However, mixers 16 may comprise any ofdouble-balanced, single-balanced, or unbalanced designs. Moreover,mixers 16 may be active or passive. Summing circuit 24 comprises anysuitable device or circuitry that adds signals 22 from mixers 16 to formIF signal 14. In a particular embodiment, summing of the mixer outputs22 is accomplished by wire-ORing the collectors of the Gilbert celloutputs. The weighting factors 20 can be applied to mixers 16 using theemitter load in the g_(m) section of the Gilbert cells.

Phase generation circuit 26 comprises any suitable combination andarrangement of devices used to generate the phase signals 18 describedherein. Examples of phase generation circuit 26 are provided in laterFIGURES. In general, a voltage controlled oscillator 28 comprises anoscillator where a control voltage controls the oscillator outputfrequency. VCO 28 can be built using many circuit techniques. In oneembodiment, the buffered output of VCO 28 is used to drive phasegeneration circuit 26 and, ultimately, mixers 16. In order to preciselytune and stabilize VCO 28, a phase lock loop (PLL) maybe used to lockthe VCO 28 to a multiple of a reference frequency provided by a crystaloscillator. For television system applications, a reference frequency of4 MHz may be used.

FIG. 3 illustrates the IF signal 14, V_(o), of circuit 10 versus timewith the RF signal 12, V_(i), held at one. With appropriate weightingfactors, w_(i), applied to mixers 16, the IF signal 14 contains nothird, fifth or even harmonics. In general, any number of harmonics canbe suppressed by increasing the number of mixers 16 and associated phasesignals 18. For N mixers numbered 0, 1, 2 . . . N−1, the phase signal 18of mixer i is given by the following equation:

${{\phi_{i}(t)} = {{{\phi_{0\;}\left( {t - {\frac{T}{2\; N}i}} \right)}\mspace{14mu}{for}\mspace{14mu} i} = 1}},{{2\mspace{14mu}\ldots\mspace{14mu} N} - 1}$where φ₀(t) is a ±1 square wave at the local oscillator fundamentalfrequency, f_(LO).

If one cycle of the desired, sampled phase signal is given by thefollowing equation:

${{v_{LO}(k)} = {{{\cos\left( {\pi\frac{k}{N}} \right)}\mspace{14mu}{for}\mspace{14mu} k} = 0}},1,{{2\mspace{14mu}\ldots\mspace{14mu} 2\; N} - 1}$Then it can be shown that the weighting factors, w_(i), are given by thefollowing equation:

${w_{i} = {{{\sin\left( {\frac{\pi}{2\; N}\left( {{2\; i} - 1} \right)} \right)}{\sin\left( \frac{\pi}{2\; N} \right)}\mspace{14mu}{for}\mspace{14mu} i} = 0}},1,{{2\mspace{14mu}\ldots\mspace{14mu} N} - 1.}$Ignoring signs, this results in N/2 unique weighting factors, w_(i).

According to a second embodiment, the phase signals are sampledaccording to the following equation:

${{v_{LO}(k)} = {{{\cos\left( {\pi\frac{k + 0.5}{N}} \right)}\mspace{14mu}{for}\mspace{14mu} k} = 0}},1,{{2\mspace{14mu}\ldots\mspace{14mu} 2\; N} - 1}$In this case, the weighting factors, w_(i), are given by the followingequation:

${w_{i} = {{{\sin\left( {\frac{\pi}{N}i} \right)}{\sin\left( \frac{\pi}{2\; N} \right)}\mspace{14mu}{for}\mspace{14mu} i} = 0}},1,{{2\mspace{14mu}\ldots\mspace{14mu} N} - 1.}$This also results in N/2 unique weighting factors, w_(i), but eliminatesone mixer 16 since w_(o) is zero.

For I-Q frequency translation, the quadrature mixers 16 are identical tothe in-phase mixers 16 illustrated in FIG. 1 except that the phasesignals 18 applied to the quadrature mixers 16 are shifted by ninetydegrees and/or inverted. To simplify the ninety degree shift, N may bechosen to be an even number. In this case, the spectrum of theequivalent complex phase signals is zero except at frequencies (2Nm+1)f_(LO) where m is any integer. The first problem spur (lowest frequency)occurs at (1−2N) f_(LO).

A frequency translation receiver, such as a direct-down conversionreceiver, is a primary application for harmonic suppression mixing. Inorder to illustrate its application, an example is presented. Assumethat the direct-down conversion receiver is designed to tune to an RFsignal 12 having signals between from 57 to 849 MHz (e.g., cabletelevision signals) and that no RF energy exists above 852 MHz. Oneapproach is to implement the direct-down conversion receiver with a pair(I and Q) of 8-phase (e.g., N=8 for a total of sixteen mixers 16)harmonic suppression mixers 16. The mixer pair will produce nounsuppressed spurious responses to any frequencies within the band.Consider the most demanding requirement. At the lowest tuned frequency(f_(LO)=57 MHz), the first unsuppressed spurious response occurs at(1−2N) f_(LO)=−15, f_(LO)=855 MHz. Since this response is greater thanthe highest in-band frequency (852 MHz), an 8-phase mixer pair isadequate for this application.

For the best harmonic rejection (and I/Q quadrature), the phase signals18 should be generated using synchronously clocked (using the VCO as theclock) digital logic. This means that the highest VCO frequency is 2Nf_(LO)=2*8*849 MHz=13.584 GHz. A technique to lower the VCO frequencywould increase the usefulness of system 10.

FIG. 4 illustrates a table 50 that shows how the example design can bebroken into four frequency bands where all but the first band span anoctave. Although FIG. 4 is illustrated and the remaining FIGURES aredescribed with reference to four bands of RF signal 12, it should beunderstood that RF signal 12 may be associated with any suitable numberand arrangement of radio frequency bands according to particular needsor desires. Table 50 comprises columns 52-64. Columns 52 and 54 identifythe band and corresponding tuned frequency range associated with thesignal of interest. Column 56 identifies the number, N, of distinctphase signals 18 used for each band and tuned frequency range. Column 58identifies the VCO division factor, M, used to generate φ₀. Theremaining phase signals 18 are generated by delaying φ₀ with delayelements, such as D-flip-flops, as explained in detail below. Column 60illustrates the lowest harmonic that is not suppressed by the circuit10. This is also referred to as the first spur harmonic. Column 62illustrates the worst case spur frequency, i.e. the frequency of thelowest unsuppressed spur when tuned to the low end of the band. Column64 illustrates the VCO tuning range used to generate phase signals 18for any given band.

Referring to FIG. 5, the first technique is illustrated by circuit 100that comprises switching circuit 102 coupled to a plurality of mixerstages 104 a-c. Mixer stages 104 a-c are referred to collectively asmixer stages 104 and generically as mixer stage 104. Each mixer stage104 is configured to work with a particular range of frequencies, orbands, of RF signal 12. For example, mixer stage 104 a is associatedwith bands 1 and 2 of RF signal 12. Mixer stage 104 b is associated withband 3 of RF signal 12. Mixer stage 104 c is associated with band 4 ofRF signal 12. Each mixer stage 104 and its configuration and operationis described in greater detail with reference to FIGS. 6-14.

Switching circuit 102 may be implemented using any suitable number,combination, and arrangement of digital and analog switching techniques,and is depicted as a series of mechanical switches for illustrativepurposes only. In the embodiment depicted in FIG. 5, switching circuit102 communicates RF signal 12 to a selected one of the plurality ofmixer stages 104 in response to a control signal 106. The control signal106 may be generated by other parts of a direct down-conversionreceiver, for example, such as by a tuner. Control signal 106 mayinstruct switching circuit 102 to communicate RF signal 12 to aparticular mixer stage 104 or may include information about thefrequency or band of frequencies associated with the signal of interest,and switching circuit 102 may determine thereupon the appropriate mixerstage 104 to which to communicate RF signal 12. By communicating RFsignal 12 to a selected mixer stage 104 according to the frequency bandwithin which the signal of interest resides, circuit 100 ensures thatappropriate harmonic suppression mixers and phase generation logic areused to frequency translate the RF signal 12 to an IF signal 14.

FIG. 6 illustrates one embodiment of mixer stage 104 c that includesmixers 16 and phase generation circuit 26 that generates phase signals18 in response to VCO signal 30 from VCO 28. Mixer stage 104 c isconfigured to frequency translate an RF signal 12 having a signal ofinterest in band 4 (e.g., 424.5-849 MHz). According to table 50 of FIG.4, the number of phases, N, for band 4 is two (e.g., φ₀ and φ₂). Eachmixer 16 of mixer stage 104 c combines the RF signal 12 with theappropriate phase signal 18 to form at least a portion of IF signal 14.IF signal 14 comprises a real part, I, and an imaginary part, Q.

FIG. 7 illustrates one embodiment of phase generation circuit 26 used inmixer stage 104 c. Phase generation circuit 26 comprises a frequencydivider circuit 110 coupled to a delay circuit 112. Frequency dividercircuit 110, also referred to as a “divide-by-N” circuit, divides thefrequency of incoming VCO signal 30 by a division factor, M. Frequencydivider 110 of FIG. 7 has a division factor, M, of four, as illustratedin table 50 for band 4. Delay circuit 112 may comprise a flip-flopcircuit, such as a D-flip-flop circuit having D and CLK inputs and Q and−Q outputs. Phase generation circuit 26 of FIG. 7 generates phasesignals 18 (e.g., φ₀ and φ₂) illustrated in FIG. 8.

FIG. 9 illustrates one embodiment of mixer stage 104 b that includesmixers 16 and phase generation circuit 26 that generates phase signals18 in response to VCO signal 30 from VCO 28. Mixer stage 104 b isconfigured to frequency translate an RF signal 12 having a signal ofinterest in band 3 (e.g., 212.25-424.5 MHz). According to table 50 ofFIG. 4, the number of phases, N, for band 3 is four (e.g., φ₀, φ₁, φ₂and φ₃). Each mixer 16 of mixer stage 104 b combines the RF signal 12with the appropriate phase signal 18 to form at least a portion of IFsignal 14. An appropriate weighting factor, w_(i), is also applied to RFsignal 12 before it is received by each mixer 16. The weighting factors,w_(i), of the second set of mixers 16 (e.g., used to generate Q part ofIF signal 14) are shifted and/or inverted with respect to the weightingfactors, w_(i), of the first set of mixers 16 (e.g., used to generatethe I part of IF signal 14). This is done in order to appropriatelyshift the phase signals 18 applied to the second set of mixers 16 byninety degrees. Summing circuits 24 combine the outputs 22 of mixers 16to generate IF signal 14 having a real part, I, and an imaginary part,Q.

FIG. 10 illustrates one embodiment of phase generation circuit 26 usedin mixer stage 104 b. Phase generation circuit 26 comprises a frequencydivider circuit 110 coupled to a plurality of delay circuits 112.Frequency divider 110 of FIG. 10 has a division factor, M, of eight, asillustrated in table 70 for band 3. Phase generation circuit 26 of FIG.10 generates phase signals 18 (e.g., φ₀, φ₁, φ₂ and φ₃) illustrated inFIG. 11.

FIG. 12 illustrates one embodiment of mixer stage 104 a that includesmixers 16 and phase generation circuit 26 that generates phase signals18 in response to VCO signal 30 from VCO 28. Mixer stage 104 a isconfigured to frequency translate an RF signal 12 having a signal ofinterest in either of band 1 (e.g., 57-106.125 MHz) or band 2 (e.g.,106.125-212.25 MHz). According to table 50 of FIG. 4, the number ofphases, N, for bands 1 and 2 is eight (e.g., φ₀, φ₁, φ₂, φ₃, φ₄, φ₅, φ₆and φ₇). An appropriate weighting factor, w_(i), is also applied to RFsignal 12 before it is received by each mixer 16. The weighting factors,w_(i), of the second set of mixers 16 (e.g., used to generate Q part ofIF signal 14) are shifted and/or inverted with respect to the weightingfactors, w_(i), of the first set of mixers 16 (e.g., used to generatethe I part of IF signal 14). This is done in order to appropriatelyshift the phase signals 18 applied to the second set of mixers 16 byninety degrees. Summing circuits 24 combine the outputs 22 of mixers 16to generate IF signal 14 having a real part, I, and an imaginary part,Q.

FIG. 13 illustrates one embodiment of phase generation circuit 26 usedin mixer stage 104 a. Phase generation circuit 26 comprises at least onefrequency divider circuit 110 coupled to a plurality of delay circuits112. When used for band 2, frequency divider 110 of FIG. 13 has adivision factor, M, of sixteen, as illustrated in table 70 for band 2.Phase generation circuit 26 of FIG. 13 generates phase signals 18 (e.g.,φ₀, φ₁, φ₂, φ₃, φ₄, φ₅, φ₆ and φ₇) illustrated in FIG. 14 used for band2. When used for band 1, another frequency divider circuit 110 having adivision factor, M, of 2, is switched into communication with frequencydivider circuit 100 having a division factor, M, of sixteen to create aneffective division factor of thirty-two, as illustrated in table 70 forband 1. Phase generation circuit 26 of FIG. 13 generates phase signals18 (e.g., φ₀, φ₁, φ₂, φ₃, φ₄, φ₅, φ₆ and φ₇) illustrated in FIG. 14 usedfor band 1.

FIG. 15 illustrates one embodiment of phase generation circuit 26 usedto generate phase signals 18 for use with mixers 16. Phase generationcircuit 26 generates a phase output based on n inputs using a decoder200. Examples of decoders 200 include the circuits used in DirectDigital Synthesizers (DDSs) and flash digital-to-analog converters. Suchcircuits are sometimes referred to as “thermometer decoders.” Decoder200 may be represented as an n-to-N truth table that maps n binaryinputs 204 to N binary outputs (phase signals 18). An example of a truthtable 202 for a 4-to-8 decoder 200 is illustrated in FIG. 16.

Truth table 202 illustrates outputs (phase signals 18) produced by4-to-8 decoder 200 in response to inputs 204. Inputs 204 act as bits ofa digital number i, shown at the left of table 202, that ranges from 0to (2^(n)−1). This number i may in turn be thought of as an input angleθ, where θ=2π(i/2^(n)), such that phase signals 18 output by decoder 200represent the phase corresponding to the particular input angle. Thus,by changing the values of inputs 204 with a certain frequency, decoder200 is able to produce a desired phase output.

In the depicted embodiment, decoder 200 is controlled by a phaseregister 206, which is in turn controlled by VCO 28. Phase register 206is an n-bit digital memory with an input 208 and an output that servesas input 204 to decoder 200. When triggered by a clock signal 210 fromVCO 28, phase register 206 reads an n-bit number at input 208 andoutputs this n-bit number as input 204 to decoder 200. Since phaseregister 206 serves as input 204 to decoder 200, the phase signals 18may be controlled by controlling the contents of phase register 206.

Input 208 to phase register 206 is generated by adder 212. Adder 212 isany digital circuit for adding binary numbers to produce an n-bitdigital output. In the depicted embodiment, adder 212 has two inputs.The first input of adder 212 is from a frequency register 214. Frequencyregister 214 is a digital memory that stores an amount by which phaseregister 206 is to be incremented when triggered by clock signal 210.The second input of adder 212 is the output of phase register 206. Thus,each time phase register 206 is triggered, adder 212 produces an outputequal to the previous output of phase register plus the incrementspecified in frequency register 214. When phase register 206 istriggered the next time, phase register 206 replaces its contents withthe output of adder 212, which effectively increments phase register 206by the amount stored frequency register 214.

The rate at which phase register 206 is triggered by VCO 28 may beadjusted using a prescaler 216. Prescaler 216 divides the frequency ofVCO 28 by a predetermined amount. The amount by which VCO 28 is dividedmay be determined based on the desired frequency band for which phasegeneration circuit 26 is intended to produce particular phase signals18. Depending on the amount M by which prescaler 216 divides thefrequency of VCO 28 and the amount by which frequency register 214increments phase register 206, a desired multiple of the frequency ofVCO 28 may be selected. This multiple may be adjusted so that the outputproduces by phase generation circuit 18 falls within a desired frequencyband.

In operation, a frequency band of interest is selected by adjusting thevalues of prescaler 216 and frequency register 214 to produce a desiredrate by which phase register 206 is incremented. Because of prescaler216, phase register 206 will be incremented every time VCO 28 completesM cycles. Each time phase register 206 is triggered, it will beincremented by the amount specified in frequency register 214. Thisdetermines which phase signals 18 will be produced by decoder 200 andhow often they will be produced. Thus, phase generation circuit 26allows band selection based on the frequency of VCO 28.

FIG. 17 represents another embodiment of phase generation circuit 26using decoder 200. In the depicted embodiment, phase register 206 is acounter that maintains an n-bit value. Phase register 206 is incrementedby a fixed amount every time phase register 206 is triggered by clocksignal 210 from prescaler 216 coupled to VCO 28. For example, phaseregister 206 may be incremented by one every M cycles of VCO 28. Unlikethe embodiment depicted in FIG. 15, phase register 206 does not read anew value every time it is triggered, but the output signal produced byphase register 206 is still the n-bit value stored in phase register206. Selected output bits of phase register 206 are coupled to AND gates218. Each AND gate 218 is controlled by a respective band selector 220.Band selector 220 is any suitable control circuitry that is used to turnthe output of AND gate 218 on and off. This effectively allows certainbits of the output of phase register 206 provided to decoder 200 to beset to zero, regardless of the actual output of phase register 206. Bandselector 220 may also be used to control prescaler 216, such as, forexample, to set the multiplier of prescaler 216 or to turn prescaler 216on and off.

In operation, a frequency band of interest is selected by setting thevalues of prescaler 216 and band selectors 220. Prescaler 216 controlsthe rate at which phase register 206 is triggered by VCO 28. Each timephase register 206 is triggered, phase register 206 increments andoutputs the incremented value. However, depending on which bits aresuppressed by AND gates 218, input bits 204 provided to decoder 200 maybe different than the output of phase register 206. For example, ANDgates 218 may replace the two least significant bits of the output ofphase register 206 with zeroes, so that only the unsuppressed bitsaffect the output of decoder 200. Thus, setting band selectors 220 alsosets which phase signals 18 are produced by decoder 200 and the rate atwhich they are produced. Again, the net result is to allow decoder 200to transition through various combinations of phase signals 18 at a ratethat is a desired multiple of the frequency of VCO 28. An example of theoutput for 4-by-8 decoder 200 with the topmost AND gate 218 turned offand the next lowest AND gate 218 turned on is depicted in FIG. 18.

Although embodiments of the invention and their advantages are describedin detail, a person skilled in the art could make various alterations,additions, and omissions without departing from the spirit and scope ofthe present invention as defined by the appended claims.

What is claimed is:
 1. A phase generation circuit, comprising: a decoder operable to: receive a plurality of input bits of an n-bit value; and generate a plurality of phase signals in response to receiving the input bits; a phase register coupled to the decoder and operable to store the n-bit value and further operable to output the n-bit value to the decoder in response to a triggering signal; a control circuit that is operable to cause one or more selected bits of the n-bit value output to the decoder to be set to zero regardless of the actual-value of the bit output of the phase register; a voltage controlled oscillator operable to generate a voltage signal at a frequency, wherein the triggering signal for the phase register is generated at least in part based on the frequency of the voltage controlled oscillator; and a prescaler operable to divide the frequency of the voltage signal by an integer to control the rate at which the phase register is triggered by the voltage controlled oscillator, wherein the control circuit performs frequency band selection by setting the one or more selected bits of the n-bit value to zero and thereby determining which phase signals are generated by the decoder.
 2. The phase generation circuit of claim 1, wherein the control circuit is further operable to set the value of the integer of the prescaler.
 3. A phase generation circuit, comprising: a decoder operable to: receive a plurality of input bits of an n-bit value; and generate a plurality of phase signals in response to receiving the input bits; a phase register coupled to the decoder and operable to store the n-bit value and further operable to output the n-bit value to the decoder in response to a triggering signal; a control circuit that is operable to cause one or more selected bits of the n-bit value output to the decoder to be set to zero regardless of the actual-value of the bit output of the phase register; a voltage controlled oscillator operable to generate a voltage signal at a frequency, wherein the triggering signal for the phase register is generated at least in part based on the frequency of the voltage controlled oscillator; and a prescaler operable to divide the frequency of the voltage signal by an integer to control the rate at which the phase register is triggered by the voltage controlled oscillator, wherein the n-bit value is incremented by a predetermined amount that is determined at least in part based upon a comparison of a first frequency range of the voltage controlled oscillator and a second frequency range comprising a signal of interest.
 4. The phase generation circuit of claim 3, wherein: the first frequency range comprises 1698 MHz to 3396 MHz; and the second frequency range comprises one of four bands in a radio frequency range, the bands comprising: 57 MHz to 106.125 MHz; 106.125 MHz to 212.250 MHz; 212.250 MHz to 424.5 MHz; and 424.5 MHz to 849 MHz.
 5. A method for generating phase signals, comprising: storing an n-bit value; outputting the n-bit value based at least in part on a voltage signal; modifying one or more selected bits of the output n-bit value such that the one or more selected bits are set to zero regardless of the actual value or values of the one or more selected bits; generating phase signals in response to the modified n-bit value; incrementing the stored n-bit value; repeating the outputting, modifying, and generating steps after n-bit value is incremented; and selecting a frequency band by setting the one or more selected bits of the n-bit value to zero and thereby determining which phase signals are generated.
 6. The method of claim 5, further comprising dividing the frequency of the voltage signal by an integer to control the rate at which the n-bit value is output.
 7. A method for generating phase signals, comprising: storing an n-bit value; outputting the n-bit value based at least in part on a voltage signal; modifying one or more selected bits of the output n-bit value such that the one or more selected bits are set to zero regardless of the actual value or values of the one or more selected bits; generating phase signals in response to the modified n-bit value; incrementing the stored n-bit value; and repeating the outputting, modifying, and generating steps after n-bit value is incremented, wherein the n-bit value is incremented by a predetermined amount that is determined at least in part based upon a comparison of a first frequency range of a voltage controlled oscillator and a second frequency range comprising a signal of interest.
 8. The method of claim 7, wherein: the first frequency range comprises 1698 MHz to 3396 MHz; and the second frequency range comprises one of four bands in a radio frequency range, the bands comprising: 57 MHz to 106.125 MHz; 106.125 MHz to 212.250 MHz; 212.250 MHz to 424.5 MHz; and 424.5 MHz to 849 MHz.
 9. A circuit for frequency translating a radio frequency signal, comprising: a decoder operable to: receive a plurality of input bits; and generate a plurality of phase signals in response to receiving the input bits; a phase register coupled to the decoder operable to: store a binary number; increment the binary number by a predetermined amount; and output the binary number to the decoder in response to a triggering signal, wherein the input bits of the decoder are based at least in part on the binary number output by the phase register; a voltage controlled oscillator operable to generate a voltage signal at a frequency, wherein the triggering signal for the phase register is generated at least in part based on the frequency of the voltage oscillator; and a plurality of mixers coupled to the decoder, each mixer operable to receive one of the phase signals from the decoder and each mixer further operable to combine the respective phase signal weighted with a respective weighting factor for the mixer with a radio frequency signal; an adder coupled to an input and an output of the phase register, the adder operable to: receive the binary number from the phase register; add the predetermined amount to the binary number to generate a new binary number; and provide the new binary number to the phase register, wherein the phase register increments by reading the new binary number from the adder.
 10. A circuit for frequency translating a radio frequency signal, comprising: a decoder operable to: receive a plurality of input bits; and generate a plurality of phase signals in response to receiving the input bits; a phase register coupled to the decoder operable to: store a binary number; increment the binary number by a predetermined amount; and output the binary number to the decoder in response to a triggering signal, wherein the input bits of the decoder are based at least in part on the binary number output by the phase register; a voltage controlled oscillator operable to generate a voltage signal at a frequency, wherein the triggering signal for the phase register is generated at least in part based on the frequency of the voltage oscillator; and a plurality of mixers coupled to the decoder, each mixer operable to receive one of the phase signals from the decoder and each mixer further operable to combine the respective phase signal weighted with a respective weighting factor for the mixer with a radio frequency signal; and a gate coupled to at least one output bit of the phase register, the gate operable to replace the value of the output bit with a predetermined bit value, wherein the input bits to the decoder comprise the replaced output bit along with the remaining bits of the binary number. 